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74017 データシート(PDF) 7 Page - Skyworks Solutions Inc. |
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74017 データシート(HTML) 7 Page - Skyworks Solutions Inc. |
7 / 10 page On the Direct Conversion Receiver CX74017 101735A Skyworks Solutions, Inc., Proprietary and Confidential 7 July 20, 2001 Preliminary Data Subject to Change It can be seen from these equations and in Figure 13 that the DC component due to the second order non-linearity is growing with twice the slope of the fundamental on a logarithmic scale. At the intercept point, IIP2 a a A A a A a = = ⇔ = 2 1 1 2 2 2 2 Due to the doubled slope of the second-order product, IM2 Pout with Pin IIP2 − = ∆ ∆ + = Noise Low frequency noise becomes a great concern in DCR [14], as significant gain is allocated to baseband stages after the mixer. Weak baseband signal levels of a few millivolts are still very vulnerable to noise. This requires stronger RF stage gain to alleviate the poor noise figure of baseband blocks, but of course this must be traded against the linearity problems, just described, that accompany higher RF gain. Flicker noise, that is, 1/f noise, is the major baseband noise contributor. Associated with DC flow, it has a spectral response proportional to 1/f. In RF circuits, 1/f noise tends to be modulated onto the RF signal. In the case of a mixer with baseband output, 1/f noise sees especially high conversion gain. In practice, flicker noise becomes an issue for Metal Oxide Semiconductor (MOS) devices more than bipolar, and is modeled as a voltage source in series with the gate. 1/f noise complicates the use of MOS transistors for RF circuits, since the main method of reducing it in MOS is to increase the transistor’s size, which increases the device capacitance. This adversely affects RF gain. For this reason, it is preferable to use bipolar transistors for DCR mixer designs. In the first baseband stages after the mixer, it becomes possible to use MOS devices, as the transistor-size tradeoff is feasible at low frequencies. I/Q mismatches Due to the high frequency of the LO, it is not possible to implement the IQ demodulator digitally. An analog IQ demodulator exhibits gain and phase imbalances between the two branches, as well as the introduction of DC offsets. Such imperfections distort the recovered constellation. Assuming ϕ α, being the amplitude and phase mismatch respectively between the demodulator quadrature ports, and the complex signal incident upon it having in-phase and quadrature components I and Q: () () ( ) ) sin( 1 2 ) sin( ) cos( ) cos( 2 ) sin( ) cos( ϕ ω α ω ω ω ω ω + + ⋅ + = ⋅ + = t t Q t I Qout t t Q t I Iout Filtering out the high frequency terms: ()( )) cos( ) sin( 1 ϕ ϕ α Q I Qout I Iout + − + = = 101735A 13_071801 1 2 1 1 Pin Pout fu nd am en tal IP2 IIP2 OIP2 IP3 1 3 1 2 1 1 Pin Pout fu nd am en tal IP2 IIP2 +49dBm OIP2 Pin Pout ∆ ∆ Figure 13. Second Order Intercept Point (IP2) |
同様の部品番号 - 74017 |
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同様の説明 - 74017 |
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