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AD8170AR データシート(PDF) 9 Page - Analog Devices |
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AD8170AR データシート(HTML) 9 Page - Analog Devices |
9 / 16 page AD8170/AD8174 –9– REV. 0 Equation 4 can be used to calculate expected gain error due to the current feedback amplifier’s finite transimpedance and common mode rejection. For low gains and recommended feedback resistors, this will be typically less than 0.4%. For most applications with gain greater than 1, the dominant source of gain error will most likely be the ratio-match of the external resistors. All of the dominant contributors to gain error are associated with the buffer amplifier and external resistors. These do not change as different channels are selected, so channel-to-channel gain match of less than 0.05% is easily attained. G = 1+ RF RG RT RT + RIN 1+ RF RG + RF 1 − CMRR [] (4) ↑ ↑ Ideal Gain Error Terms RT = Amplifier Transresistance = 600 k Ω RIN = Amplifier Input Resistance ≅ 100 Ω CMRR = Amplifier Common-Mode Rejection ≅ –52 dB Choice of External Resistors The gain and bandwidth of the multiplexer are determined by the closed-loop gain and bandwidth of the onboard current feedback amplifier. These both may be customized by the external resistor feedback network. Table III shows typical bandwidths at some common closed loop gains for given feedback and gain resistors (RF and RG, respectively). The choice of RF is not critical unless the widest and flattest frequency response must be maintained. The resistors recom- mended in the table result in the widest 0.1 dB bandwidth with the least peaking. 1% resistors are recommended for applications requiring the best control of bandwidth. Packaging parasitics vary between DIP and SOIC packages, which may result in a slightly different resistor value for optimum frequency performance. Wider bandwidths than those listed in the table can be attained by reducing RF at the expense of increased peaking. To estimate the –3 dB bandwidth for feedback resistors not listed in Table III, the following single-pole model for the current feedback amplifier may be used: ACL = G 1 + sC T RF +GN RIN () ACL = Closed Loop Gain CT = Transcapacitance 0.8 pF RF = Feedback Resistor G = Ideal Closed Loop Gain GN = (1 + RF/RG) = Noise Gain RIN = Inverting Terminal Input Resistance ≅ 100 Ω The –3 dB bandwidth is determined from this model as: f –3dB ≅ 1 2 π C T RF +GN RIN () This model is typically good to within 15%. Table III. Recommended Component Values Small Signal Large Signal VOUT = 50 mV rms VOUT = 0.707 V rms Gain RF ( )RG ( ) –3 dB BW (MHz) –3 dB BW (MHz) AD8170R +1 1 k — 710 270 +2 499 499 250 290 +10 499 54.9 50 55 +20 499 26.3 27 27 AD8174R +1 1 k — 780 270 +2 549 549 235 280 +10 499 54.9 50 55 +20 499 26.3 27 27 Capacitive Load The general rule for current feedback amplifiers is that the higher the load capacitance, the higher the feedback resistor required for stable operation. For the best combination of wide bandwidth and clean pulse response, a small output resistor is also recommended, as shown in Figure 24. Table IV contains values of feedback and series resistors that result in the best pulse response for a given load capacitance. RG VIN SWITCH RF RT 50 Ω VOUT 0.1µF 10µF BUFFER +VS 0.1µF 10µF –VS RS(OUT) CL (TO FET PROBE) Figure 24. Circuit for Driving a Capacitive Load Table IV. Recommended Feedback and Series Resistors and Bandwidth vs. Capacitive Load and Gain G = +1 G = +2 G = +3 G +4 VOUT = 2 V p-p VOUT = 2 V p-p VOUT = 2 V p-p CL RF RSOUT –3 dB BW RF RSOUT –3 dB BW RF RSOUT –3 dB BW RF RSOUT (pF) ( )( ) (MHz) ( )( ) (MHz) ( )( ) (MHz) ( )( ) 20 1 k 50 149 1 k 20 174 499 25 170 499 20 50 1 k 30 104 1 k 15 117 1 k 15 98 499 20 100 2k 20 73 1 k 15 80 1 k 15 71 499 15 300 2k 20 27 1 k 15 34 1 k 15 33 499 15 |
同様の部品番号 - AD8170AR |
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同様の説明 - AD8170AR |
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